A $1.50 soil moisture sensor: ready to deploy. My first experiments with the pulsed output hack didn’t quite work due to probe polarization at low frequencies…but I’m still noodling with it.

There’s something of a cultural divide between the OpAmp/555 crowd and Arduino users. I think that goes some way to explaining the number of crummy sensor modules in the hobbyist market: engineers probably assume that anyone playing in the Arduino sandbox can’t handle anything beyond analogRead().  So it’s not unusual to find cheap IC sensor modules with cool features simply grounded out, because how many ‘duino users could config those registers anyway – right?

Legit suppliers like Adafruit do a much better job in that regard, but our project rarely uses their sensors because they are often festooned with regulators, level shifters, and other ‘user-friendly’ elements that push the sleep current out of our power budget. Sparkfun boards usually have leaner trim, but with all the cheap PCB services available now I’m tempted to just roll my own. Thing is, I keep running into the problem that in ‘prototype quantities’ the individual sub-components often cost more than complete modules with the reflow already done – and that’s before everything gets sand-bagged with shipping charges larger than the rest of the project combined.

So we still use a lot of eBay modules – after removing the usual load of redundant pullups and those ubiquitous 662k regulators.  Once you go down that rabbit hole you discover a surprising number of those cheap boards can be improved with ‘other’ alterations. The reward can be substantial, with our low power RTC mod bringing $1 DS3231 modules from ~0.1 mA down to less than 3µA sleep current – which is quite useful if you want to power the entire logger from a coin cell.

Another easily modified board is the soil moisture probe I flagged in the electrical conductivity post:

Schematic before conversion:  (regulator not shown). The output frequency is controlled by the time constant when  charging/discharging C3 through R2/R3. Gadget Reboot gives a good overview of how the basic configuration works, feeding the RC filtered 555 output into a simple peak detector. Older NE555 based probes run at 370khz but the V1.2 probes with the TLC555 run at a higher 1.5Mhz frequency with a 34% duty cycle. It’s worth noting that dissolved salts, etc affect soil moisture readings until you reach the 20-30MHz range.

These sensors use coplanar traces to filter high frequency output from a 555 oscillator, but you’re lucky to see a range of more than 400 raw counts reading that filtered output with Arduino’s 10-bit ADC.  On 3.3v systems, you can remove the regulator and combine the sensor with an ADS1115 for 15-bit resolution. I found you can’t push that combination past the ±4.096V gain setting or the 1115’s low input impedance starts draining the output capacitor. But that still gives you a working range of several thousand counts as long as you remember that the ADS1115 uses an internal vRef – so you also need to monitor the voltage supplied to the sensor to correct for battery variation.  Trying to read the sensors output with my EX330 when the sensor was running brought the output voltage down by about 0.33v and it’s got 10MΩ internal impedance similar to the ADS – so a cheaper voltmeter will knock the output from this sensor around even more. Differential readings can reduce this problem because that doubles the ADC’s impedance.

Analog mode works well with long cable runs and piping that through an ADS1115 lets you communicate with three soil sensors over I2C if you are low on ports.  And up to four of those $1 ADC could be hung off the same bus – though I’ve learned the hard way not to put too many sensors on one logger because it risks more data loss if you have a point failure due to battery leaks, critters, or vandalism. It only takes a couple of hours to build another logger for each set anyway.

3.3v ProMini ADC readings from an analog soil moisture sensor at ~8cm depth (vertical insertion) with  DS18b20 temp from the same depth. We ran several of these sensors in our back yard this summer. This segment shows daily drawdown by vegetation during the hottest/driest month of the year – followed by several rainfall events starting on 9/4 . This logger was running unregulated from 2x lithium AA cells and the regulator was also removed from the soil sensor. Aref moves in step with the supply voltage so we didn’t have interference from battery variation. C5&6 were removed from the board and the sensor was powered from a digital pin with 8 seconds to stabilize ( probably more time than needed). This humble sensor was less affected by daily temperature cycles than I was expecting but if you use ‘field capacity’ as your starting point the delta was only ~200 raw ADC counts. I manually watered the garden on 8/23 & 8/24 to keep the flowers from dying; but it was only a brief sprinkle. We have 6 months of continuous operation on these probes so far but with only that thin ink-mask protecting the copper traces I’d be surprised if they go more than a year. Even with epoxy encapsulating the 555 circuit, small rocks could easily scratch the probe surface on insertion leading to accelerated corrosion.

These things are cheap enough that I’ve started noodling around with them for other tasks like measuring water level.  With a water/air dielectric difference of about 80:1 even unmodified soil probes do that job well if you put in a little calibration time. When powered at 3.3v, the default config outputs a range of ~3.0v dry to 1.5v fully submerged in water. So Arduino’s 10-bit ADC gives you decent resolution over the probes 10cm length. And since the TL555 doesn’t draw more than 5mA (after settling), you can supply it from a digital I/O pin to save power – provided you give the sensor a couple of seconds to charge the output capacitor after power on. Adding 120-150Ω in series to throttle capacitor inrush still leaves you with ~1.2v air/water delta. The sensors I tested stabilize at power on after ~1 second but take more than 35 seconds to ‘discharge’ down if you suddenly move the probe from air to water – this behavior indicates the boards are missing the R4 ground connection. (Note: After checking batches from MANY different suppliers I’ve now started adding a 1Meg ohm resistor across the output of any sensors that output ~95% of their supply voltage in ‘free air’. Sensors with the 1meg already connected the way it’s supposed to be on the board usually output ~85% of their supply voltage with the sensor in air & about 35% of their supply when completely submerged in water.)

A quick search reveals plenty of approaches to making the sensor ready for the real world, but we’ve had success hardening different types of sensor modules with a epoxy + heat-shrink method that lets you inspect the circuits over time:

Cable & Epoxy Mounting:                               (click images to enlarge)

Clean the sensor with 90% isopropyl alcohol. An old USB or phone cable works for light – duty sensor applications. Here I’ve combined two of the four wires to carry the output.	3/4″(18mm) to 1″(24mm), 2:1 heat-shrink forms a container around the circuits, with smaller adhesive lined 3:1 at the cable to provide a seal to hold the liquid epoxy.	Only fill to about 15% of the volume with epoxy. Here I’m using Loctite E30-CL. This epoxy takes 24 hours to fully cure.
GENTLE heating compresses the tubing from the bottom up. This forces the epoxy over the components	Before finishing, use a cotton swab to seal the edges of the probe with the epoxy. Cut PCBs can absorb several % water if edges are left exposed.	Complete the heating over a rubbish bin to catch the overflow & wipe the front surface of the probe so that you don’t have any epoxy on the sensing surfaces.

Soil moisture sensors measure the volumetric water content of a soil indirectly by using some other property such as electrical resistance or dielectric permittivity. The relation between the measurement and soil moisture varies depending on environmental factors such as soil type, temperature, or the amount of salt dissolved in the pore water.  This is kind of obvious if you think about how differently sand behaves with respect to water infiltration & retention compared to a soil high in clay content – but the low level details get quite complex.

I generally use our Classroom logger for backyard experiments. You want at least 2-3 monitoring locations due to the spatial variability of infiltration pathways, and then 2-3 sampling depths at each spot for a complete profile. Agricultural applications usually install two sensors, with one shallow sensor at 25-30% of the root zone depth and one deep at 65-80% of the root zone depth. But research projects install up to one sensor every 10cm through the soil profile. If you are doing that then you might want to install the sensor boards parallel to the ground surface and only power them one at a time so they don’t interfere with each other.

Vernier outlines a basic volumetric calibration procedure for soil sensors which starts by measuring soil from your target area that’s been dried in an oven for 24 hours. To get another calibration point you add an amount of water equivalent to some percentage (say 5%) of that soils dry volume and take another sensor reading after that’s had a chance to equilibrate.  To get a complete response curve you would use 5-10 jars of dry soil, adding different relative %-volumes of water to each pot from 5 to 50%. (Even in clay the %volume of water at field capacity is usually less than 50%.) Then you can interpolate any intermediate sensor readings with something like Multimap.

If the distance between interdigital coplanar electrodes is comparable to the smallest dimension of each electrode, then ‘fringe fields’ are significant.  As a rule of thumb, the field usually has a penetration distance of between 0.5-1x the distance between the centers of the electrodes.  I found several papers modeling that as one third of the width of the electrode plus the gap between electrodes: EPD ≈ (W + G)/3. In this case I estimate the useful sensing distance is less, say 3-6 millimeters from the probes surface, so you have to take care there are no air gaps near the sensor surface during the calibration, and when you deploy. Agricultural sensors are often placed in an augured hole filled with a screen filtered soil slurry to avoid air bubbles. One of the biggest challenges is that even after a good deployment, the soil dries out and ‘break away’ from contact with the surface on your sensor. This will give you grief with almost all of the soil measuring sensors on the market. (with the possible exception of neutron probes) Another issue with this method is drilling the hole necessarily cuts any moisture pulling roots away from the probe, creating offsets relative to the root-permeated soil nearby. 

The story gets more complicated with respect to growing things because the pores of a sandy soil might provide more plant available water than absorbent clay soils (see: matric potential) So it’s worth taking the time to determine the texture of your soil, and you have consider the root depth of your crop. (typically 6-24 inches)  With all the confounding factors, soil sensors are often used in the context of ‘relative’ boundaries by setting an arbitrary upper threshold for the instrument’s raw output at field capacity (~ two days after a rain event) and the lower threshold when plant wilting is observed. Soil moisture sensors also need to be placed in a location that receives a similar amount of sunlight to account for evapotranspiration. Then there’s all the factors related to your sampling technique.

Using ‘field capacity’ as our upper limit might make it possible to do a basic three -point calibration of your sensors: Dig your trench and take ‘in situ’ measurements with your probe before you disturb the rest of the soil. Extract two samples  with a tubular ‘cutter’  who’s volume is large enough to cover your sensor and weigh them (you can back-calculate the gravimetric % moisture for those in-situ readings later from the sample you dry out ).  Dry one completely in an oven and the raise the other one to field capacity by adding water and then letting it drain & stabilize for a day or so in a 100% humidity chamber. (a big zip-lock bag?)  The wet sample must be allowed to drain under gravity until water is no longer actively dripping from it. Then weigh both samples again & take readings with your sensor embedded in the wet & dry soil samples packed back into their original sampling volume. The tricky part is that you probably need to use a metal tube for cutting your sample (and for the oven drying) but you don’t want metal near a capacitance probe later on – so you would want to transfer your soil plugs into a PVC pipe with the same internal volume using some kind of plunger before the final measurements. The idea is you want to keep the relative ‘density’ of the sample similar to the original soil insitu.

The Hack:                                                               (click images to enlarge)

After testing some ‘naked’ boards ( where I had removed all the components)  I realized that the bare traces fell within a workable capacitance range for the same astable configuration the timer was already configured for:

Get the regulated boards with the CMOS TLC555 chip rather than the NE555. 3.3v operation is out-of-spec for the NE and 1/2 the NE’s I’ve tried didn’t even work in their default analog config at 5v. (& the 3.3v 662k reg is below spec if your supply falls below ~3.4v)	I remove the regulator & bridge the Vin to Vout pads. If you are going to pin-power the sensor you also need to remove the reg caps C5 (10µF) & C6 (160nF) as their inrush current could damage your MCU. Or add  ~150 limit resistor in series, but that will drop output voltage.	Note that some variants of these sensors come with the regulator already removed DO NOT BUY THESE BOARDS  – they use older NE555 chips which won’t operate at 3.3v.
Remove the T4 diode, C3 & C4 caps and R2& R3 resistors as shown.	Move the10k R1 to the R3 pads, and the 1Meg ohm R4 to the R2 pads. When the new R2 exceeds is more than 10x the new R3 resistor you approach a 50% duty cycle on the FM output.	Bridge the R1/T4 pads closest to the 555. This patches the frequency output to AOUT. Linking the other T4 pad to C3 replaces put’s the probes capacitance in its place.

There isn’t much left after the mod. It’s important to leave the C1 bypass in place but I haven’t had problems pin-powering other sensor boards with similar caps. It’s the bare minimum you can get away with. (& add a series limiter if pin powering)

It’s worth mentioning that the 555’s notorious supply-current spikes during output transitions might give you bad reads or weird voltage spikes. I tried pin powering anyway because the TLC555 is so much better than the the old NE555s which would surely destroy any IO pin used to supply it.  So I’m sailing pretty close to the wind here: it would be much wiser to just leave the original C5/6 caps in place and skip pin powering all together. Using a mosfet is the safer choice, but hey – this is a hack…right? Officially, as little as 0.1uF needs a limit resistor to protect I/O pins, but ‘unofficially’ pin-powering dodgey circuits with AVR pins is surprisingly robust.

With the probe capacitance varying from <30 pF to ~400 pF (air/water), using 1M/10K resistors on the R2&R3 pads will give you about 50 kHz output in air, and ~1.5 kHz in water. Or you could drop in your own resistor combination to tune the output.  Our ProMini based loggers have a relatively slow 8MHz clock so the upper limit for measuring period intervals is about 120kHz before things get flakey. On a 16MHz board you can reach 200 kHz easily and with a faster MCU you could tweak that to even higher speeds. But at first bounce, with parts I already had, the 1M&10K combination seemed  workable.

the TLC555 draws far less current and works at 3.3v.  It can also be pushed to 2 MHz in astable mode, though you’d need a more advanced counter to keep up. LMC555 & TLC551 work at even lower voltages than TLC555

You have variety of options to read pulsed signals with an Arduino. We reviewed those in the ‘Adding Sensors to an Arduino Datalogger‘ but the short version of that is:  Tillart’s TSL235R example works, or you could try PJRC’s FreqCount Library. Gate interval methods count many output cycles. This reduces error and lets you approach frequencies to about 1/2 your uC clock speed (depending on the interrupt handling). However at lower frequencies the precision suffers, so measuring the elapsed time during a single cycle is better.  Since I’ve already switched over to using the Input Capture Unit to read thermistors, I started with  Reply #12, on Nick’s Timers & Counters page.  This works with these hacked soil sensors, though it will occasionally throw a spurious High/Low outlier with those rounded trapezoidal pulses. Putting a few repeats through Paul Badgers digital smooth filter crops away those glitch reads and median filters are also good for single spike suppression.  Denes Sene’s Simple Kalman filter is also fun to play with, though in this case that’s  killing flies with a sledge hammer.

As usual, I pretty much ignored all the calibration homework and simply stuck the thing in the ground for a couple of weeks to see what I’d get. Which was fortunate in a way, because the sensor developed a problem which I probably would have missed during short calibration runs:

Probe Frequency [Hz] (left axis) at 5cm soil depth. Rain event on 9/5 reset the probe to normal for a few days.

The  daily thermal wobble was expected, but the probe repeatedly displayed an upward bend which did not match the daily ‘stair-step’ drying cycles seen with other nearby soil sensors. Were we slowly pulling the ions out of the soil matrix? Rain events would reset back to normal  behavior, but eventually the rising curve would happen again.  Perhaps that was because the water was leaving, but the ions were being held on the probe surface?  When I looked into the frequency of commercial sensors, they were running at least 100 kHz, and most were in the 10’s of MHz range. Some of them automatically increased frequency with increasing conductance specifically to avoid this type of problem.

While the heat shrink/epoxy method is our gold standard for sensor encapsulation, adhesive lined 3:1 heat-shrink can do a reasonable job on these sensors if you make sure the surfaces are super-clean with isopropyl alcohol & take time to carefully push out any air bubbles. (use gloves so you don’t burn your fingers!) A third alternative is to use hot glue inside regular heat shrink tubing – squashing it into full contact with the circuits while the glue is still warm & pliable. You still have to treat the edges of the PCB, but that can also be done with nail polish. While much faster to prepare: the heat-shrink (shown in the photo above) & hot glue methods will‘pull away’ from the smooth sensor surface after about 1 year in service – epoxy encapsulation lasts for the lifetime of the probe.

I also tried to measure the electrical conductivity of solutions with this hacked probe. But without polarity reversals, polarization again became a limiting problem – and this only gets worse if you increase resistor values to resolve more detail with slower 555 pulses.  So at low freshwater conductivities, where polarization is negligible, the probe works ‘ok’ as a low-resolution EC sensor if you take your readings quickly & then de-power the probe for a long rest period. (while stirring the heck out of it..? ) However the readings ‘plateau’ as you approach 10 mS/cm – so it’s not much use in the kind of brackish coastal environments we play in.

At it’s heart, this is just a variation of standard RC rise-time methods with the 555 converting that into pulsed output.  While my attempts with the 1M&10K pair didn’t deliver much, the parrot ain’t dead yet.  Changing that R2/R3 resistor combination to boost frequencies to much higher frequencies when the probe is in dry soil might reduce that polarization problem.  Or, with a fixed capacitor instead of the probe I could try replacing a resistor with some plaster of Paris & carbon rods for a matric potential sensor with pulsed output. Essentially using the circuit as a cheap resistance to frequency converter.

And then there’s all the other capacitive sensors that I could jumper onto the C3 pads.  At that point I might as well get out the hack-saw, because I’m really just using the top section of the board as a TLC555 breakout that I can mount under epoxy. When you consider how many of us are working with 3.3v MCUs, a low voltage 555 module should have been on the market already.  And while I’m thinking about that – where are all the 3v op-amp boards, with each stage jumper-able into several ‘standard recipes’, and linkable to the next by solder pads or holes along the outer edge?  I’m envisioning an SMD quad in the middle, and a few layers of elegantly designed traces & well labeled through-holes for population with resistors & caps.  Even at eBay prices, you’d think the mark-up would have made those common as dirt a long time ago . . .

Selected Sources:

• Soil Sensor
• Gadget Reboot: Capacitive Soil Moisture Sensor Test
• Characterization of Low-Cost Capacitive Soil Moisture Sensors for IoT Networks
• Effective Calibration of Low-Cost Soil Water Content Sensors
• Calibration and Validation of a Low-Cost Capacitive Moisture Sensor…
• Soil Water Dynamics [nature]
• Methods & techniques for soil moisture monitoring [University of Wyoming]
• The soil water compendium [Edaphic Scientific]
• Methods to Monitor Soil Moisture
• Field Devices For Monitoring Soil Water Content
• Soil moisture and matric potential – an open field comparison of sensor systems
• Understanding Soil Water Content and Thresholds for Irrigation Management
• Selker and Or: Soil Hydrology and Biophysics
• Soil Moisture Sensor Installation [Decagon Devices]
• Soil Moisture Profile Probe Using a Capacitive Touch Sensor
• How to convert gravimetric soil water content to volumetric soil water [Edaphic]
• Installation of Soil Sensors for Irrigation Scheduling [& identifying bad data]
• Understanding Soil Water Content & Thresholds for Irrigation

Addendum: 2020-12-18

• The Food and Agriculture Organization of the United Nations has released its first ever global assessment of biodiversity in the underground world.
• A video about plant roots competing with each other under ground.
• And finally a cool looking antique gauge hack . It’s too bad they didn’t use a capacitance sensor with it – those two prong resistive soil sensors usually corrode within a few weeks.

The analog soil sensor from the start of the post is still chugging away, but (with the exception of a few rainy days) after leaf-fall the soil sensor has basically leveled out at ‘field capacity’

Two heavy rain events in this record. With no more evapotranspiration, the soil stays saturated between them.

So there won’t be much going on over winter but, I will be leaving most of the loggers outside anyway. We have some northern projects waiting in the wings and I want to see how well the unregulated student builds stand up to the cold. Be interesting to see if the soil sensor can withstand being frozen right into the soil.  I expect those events will register as ‘extremely dry’ due to the reduced dielectric of ice.

Addendum 2021-05-02

Lately it’s been challenging to get the 3.3v regulated versions of theses soil sensors. Vendors on eBay have started listing 3.3-5v compatibility, and even posting the schematic showing the regulated TLC555 circuit, and then shipping the ‘5v only’ NE555 sensors. This kind of bait & switch is common with low end stuff from China and the problem is your shipping charge from the US back to Shenzhen usually costs more than the items. So I just started replacing the NE555 chips myself, since TLC555’s are only about 50¢ each:

The Touchstone TS3002 timer IC could be another interesting 555 replacement option as it’s spec’d to draw only 1μA from a 1.8-V supply. A drain that low brings these soil sensors within the power budget of our 2-Part falcon tube loggers that run for a year on a coin cell.

Addendum 2023-03-18

From an electronics point of view, putting stuff in the ground isn’t really much different than putting it underwater. So people reading this post will probably some more useful information in our post about Waterproofing Electronics Projects.

Testing configuration w differential reads from a piezo triggering burst-samples with a $1 ADS1115 module.

The 16-bit ADS1115 has a programmable amplifier at the front end, with the highest gain setting providing a range of +/- 0.256 v and a resolution of about 8 micro volts.  But readers of this blog know you can already approach 14-16 bit sensitivity levels with Arduino’s ADC by oversampling with lower Arefs & scaled ranges. PA1EJO demonstrated a ADS1115 / thermistor combination which resolved 5 milli Kelvin, but we reach that resolution on our NTC’s using the ICU peripheral with no ADC at all. The beauty of time-based methods is that they scale gracefully with sensors that change over a huge range of values.  So why am I noodling around with this ADC module?

The primary attraction is this ADC has differential inputs. This is especially useful with Wheatstone bridge arrangements. A typical combination would be a two element varying bridge, and an inexpensive voltage reference like the LM4040, or the TL431. Adding the second sensor doubles the voltage swing, and the ADC’s  -32768 to +32767 raw output fits perfectly into Arduino’s 16-bit integer variables. It’s also worth noting that unlike most ADC’s – the ‘volts per bit’ via the gain settings are independent of the rail voltage supplying the chip. This means that the ADS1115 can measure it’s own supply using that internal reference without a divider. The drawback of that is I have to set full-scale as +/-4.096V on my 3.3v loggers, so the ADC only uses ~80% of the bit-range.   

Read the difference between A0 and A1 as a differential input, and read A2 as a single-ended input. That will give you an ‘almost‘ ratio-metric measurement because you record every voltage affecting the output. (although since the supply is not Aref like it would be in a regular ADC – the uncorrelated noise in the LM4040 excitation voltage ‘between’ those two readings will not get corrected)  I treat the ref. resistors as ‘perfect’ in my calculations, which forces their tempco errors into the thermistor constants during calibration.

The diodes shunt any voltages that exceed their vF, protecting the ADC from spikes beyond the +/-0.256v range at high gain. AND the leakage that Shottky’s are known for bleeds away residual charge on the piezo, preventing drift from the 1.65v bias point. All the data presented in this post used this circuit. Other diodes, or even garden variety LED’s, could also be used to clip the signal at different voltages. Most have far less leakage than Shottkys, so you might need to add a large value bleed resistor. If you do end up with an offset, an old trick is to run a very aggressive low pass filter on your readings to obtain the offset and then remove it by  subtraction.

Piezos can also be read with bridge arrangements if they are physically connected with alternating polarities, but that’s not usually the case for active sensors. I have a new project in the works where the sensor will only generate +/- 5mv, and I’d like to see if capturing signals that small is even possible with the ADS1115. To reveal where the weak points are I’ll test it with a single piezo disk reading at the highest gain and fastest sample rates. At this sensitivity a 5mv swing will only produce ~640 counts.  With my signal only covering 2% of the bit range, I’m hoping that the differential readings will compensate (?) for  noise on the rails. The data sheet warns about low input impedance (710kΩ) but I don’t think that will affect this test. Another significant limitation of the ADS1115 is that, like the Arduino driving it, no voltages below GND, or above Vcc are allowed on any input. Bridge arrangements automatically bias to the mid-point, so while the tie-in points might go ‘negative’ relative to each other, they are still positive relative to GND on the ADC.  For single sensor applications with +/- output, you need to provide that biasing with a couple of resistors.

An often overlooked feature of the ADS is the programmable comparator which can set threshold alarms on the ALRT/RDY output. Most loggers operate with fixed interval sampling, but this makes it difficult to measure things like airborne exposure peaks for chemical vapors; even with short intervals. Sensor-triggered sampling can also save battery power by letting you sleep the CPU – especially when you are monitoring environments that are only ‘active’ during certain seasons, or with rainfall events. The different comparator modes on the ADS1115 also offer some interesting possibilities for system control.

Driving the ADS1115:

This chip’s been around for a long time, so there are several libraries to choose from. And, as usual, many of them don’t support the features that a project building loggers would be most interested in. I suspect this is because wireless coms use such a prodigious amount of power that few in the IOT crowd would bother writing efficient code for chips that already auto-sleep. The Adafruit library even inserts 8ms delay statements – wasting cpu power and throttling the sample rate to 125sps. Rowberg’s I2Cdevlib does a better job with setConversionReadyPinMode() functions. But his code example only polls the I/O status, rather than using the hardware interrupts available on the same pin.

Perhaps the easiest starting point for beginners is the ADS1115 lite library.  This is a stripped down version of Adafruit’s lib. but Myers has removed the explicit delays and replaced them with a do-while loop which polls the Operational Status bit to see when the ADC has a new reading in the output registers. This minimalist approach uses only two main functions:

triggerConversion() – Sets config register bits  & then writes that register back to the sensor (which automatically starts a single-shot reading) The ADS1115 auto-sleeps after the reading.

getConversion() – A do-while loop forces the CPU to continuously check the Operational Status bit. That bit change breaks the loop, and getConversion then reads the 16-bit output register.

With this single-shot approach, a short for-loop takes burst of readings:

In single shot mode, you have to re-write the configuration register every time you want a reading:

Myers triggerConversion() function sets the config register with a common Bitwise-OR method. I’m going use this as a starting point, tweaking a few things for better readability and including my standard a 16-bit register functions so this page doesn’t scroll on forever.
(also note that in addition to my typos, wordpress inserts a ton of invisible cruft characters that will mess with the IDE – so don’t copy/paste directly from this post…)

Sensor Triggered Sampling:

Let’s use the comparator to start each burst of readings in response to me tapping the surface of the desk that the piezo sensor is resting on. Although Myers polling method doesn’t use the ADC’s ALERT/RDY output, we are already set up for triggered bursts because all the comparator control bits were zeroed with config = 0 at the start. 

COMP_MODE: 0    =>   Traditional hysteresis mode (On above Hi_thresh, Off below Lo_thresh)
COMP_POL:  0       =>   ALERT active brings the pin LOW
COMP_LAT: 0         =>   NON latching
COMP_QUE: 00     =>   ALERT after one reading above Hi_thresh  (01=2reads, 10=4reads)

With this as the starting point all you have to do to initiate comparator threshold alerts is load some non-default trigger values into the Hi_thresh & Lo_thresh registers.  Hi_thresh must be greater than Lo_thresh, and you have to use ‘2s complement’ values

Now we need a way to make the processor respond the ADC’s alert. If I wanted to use the same power wasting methods you find in most sensor libraries, I’d connect ALRT/RDY  from the ADC to any digital input pin, and poll the pin until it goes low:  

This might be OK for an IOT sensor hanging off of a wall-wart. But for logging applications a hardware interrupt based approach lets you save power by sleeping the processor until the trigger event happens:

With 200 / 300 set as thresholds, tapping the desk beside the piezo produced:

RAW ADC output vs Sample Number: Threshold triggered: A0-A1 differential, 475SPS, 16x PGA,500 samples

With readings above 2000, I was hitting the desk a bit too hard for that 5mv target range. And 475 samples-per-second is not quite fast enough to show piezo sensor behavior. Zooming in also shows that the ADC was aliasing through some background signal:

RAW ADC output vs Sample Number: ADS1115 & piezo sensor, A0-A1 differential, 475SPS, 16x PGA, 500 samples

That’s a classic ‘mains hum’ problem.  Annoying, but from a research perspective the loss of information from the start of the event was is a more of an issue: What happens if we only get one chance to record our event?

Pretriggered acquisition:

To capture infrequent events, I need to start acquiring data before the reference trigger. And since the waiting period is unknown, those readings need to go into a circular buffer that wraps around and stores each new sample over the oldest one in memory. With this approach the trigger event actually serves to stop the acquisition rather than to start it. And you want to do this gradually, so the samples in the array represent a “slice-in-time” covering the entire event.

The real trick is to sleep the main processor as much as possible during the pre-fetch period. In continuous conversion mode the ADS1115 can alert completion with an 8 msec pulse, but with only one alarm output, the ‘threshold detection’ will have to be done in software:

Setting countdown = numberOfSamples/2 centers the event in the array. (although 1/3 to 2/3 split might be better?) A tap on the desk with a pencil produced a +/- 5mv swing (~640 raw counts), and my breadboard proto circuit is picking up almost 100 counts of mains hum. 

RAW ADC output vs Sample #: 860sps, 16xPGA, diff. A0-A1, USB powered from laptop

Losing 20% of my available signal range to background cruft is a problem. Adding a $10 USB isolator, reduced that by about 1/3. But the 60 Hz signal was still distorting the shape of the waveform significantly or we’d be seeing a smoother damped harmonic..

RAW ADC output vs Sample #: 860sps, 16xPGA, diff. A0-A1, with USB isolator. Hit the desk a bit to hard on this one.

My first thought was ‘This is a well behaved, repeating signal – I’ll just subtract it out’. Unfortunately 860 SPS is not a multiple of 60Hz, so simple corrections tend to pass in & out of phase with the hum – eventually making the situation worse.  The misalignment means we’d need to do some serious math for the correction at 860SPS, so I’m probably not going to be implementing that filtering on an 8-bit processor. Alternatively I could go back to single shot sampling and use the processors internal timers to only request each new sample at some  whole number multiple of the 60 Hz mains cycles, like for example at 600 readings a second. The maximum possible would be 840, and you’d might want to add some jitter to that.

Next I tried a run from batteries only, with the nearby electrical devices all turned off. This reduced the mains hum by ~10x relative to the USB tethered operation:

RAW ADC output vs Sample #: 860sps, 16xpga, differential A0-A1, Battery powered logger

A dramatic improvement, with the ‘pre-event’ noise falling below +/- 10 counts. Most of our field deployments are in caves and this ADC looks like it has an acceptable noise floor for work in that kind of isolated environment . But the project also has a teaching component so I’d also like to use this ADC module in classroom settings.  Zooming in on that last graphs shows that working with tiny sensor signals will be challenging if we are anywhere inside a building – even if the resting state of the system looks OK:

Once the sensor is set in motion, even tiny interferences from the mains will reinforce each other before the system settles again. Even if I use internal timing control to synchronize the readings with a whole number multiple of the mains, it looks like I still won’t be able to use the ‘before’ data to fully correct the ‘after’ effects. This might be specific to way piezo sensor’s resonate, but I’ve got some homework to characterize the effect before we start building a student lab with this module.

Looking in the bright side, even with a power hungry 1284p based logger, the current draw while capturing the pre-event readings averaged less than ~450 μA for the whole system.

The path forward:

The successor to the ADS1115 is the 24-bit ADS1219 which reads up to 1000 SPS (20 Effective Bits, PGA x4). It has integrated input buffers to allow measurement of high impedance inputs & separate inputs for analog vref (true ratiometric!) and digital power/ground. This gives you more options to mitigate power supply noise, which as we’ve seen can be important for small signals. It also offers some built in 50-Hz and 60-Hz rejection, but only at slow sample rates. The ADS1115 is a delta-sigma converter so it continuously samples its inputs (oversampling @250kHz internally) which causes a transfer of charge. From the outside, this appears as a resistance which depends on the sampling rate and depends on the gain of the input stage. Higher gain for more sensitivity yields lower effective input resistance which adds in parallel to the resistance of your sensor circuit. So if you were reading the output of a voltage divider (equivalent) the ADS1115 itself would change the result. So input buffers on the ADS1219 are a welcome addition.

The low input impedance of the ADS1115 can prevent you from using the higher gain settings in differential mode  unless you add an opamp/buffer to prevent the ADC from putting to much drain on the sensors output. This is really what separates the ADS from ‘Instrumentation quality ‘ components which generally have much higher input impedances.

There are other 24-bit options in the hobbyist market like the HX711 (24-bit, PGA x32,64,128 – but only x32 on the 2nd channel? ) that is commonly sold with load cells, and I’ve seen it mentioned that the SPI HX711 works with libraries written for the ADS123x series. The ADS1232  (24 bit, fixed x128 gain) might be a easier option for dedicated bridge sensors, and they can be found on eBay for ~$7. One advantage of the ADS123x over the HX711 is that they have a switch that can shut off current to the sensor bridge when the ADC is in Standby or PowerDown mode. Of course then you have the problem that load cells take some time to warm up when power is applied, often taking several minutes to stabilize.   You occasionally see seismometer projects using the 32-bit ADS1262, which has a sensitivity >1000x better than the 1115, but with a fairly slow sample rate.

This circuit from Gadget Reboot shows one method of obtaining programmable gain control using an X9C digital potentiometer in the opamp feedback loop. See: Part1 & Part2    The DS3502 gives you an I2C bus version with the same 10k range, though I have no idea what the long term stability of these digital pots is. And 5% tolerance is a bit grim.

But this little experiment has me wondering if for signals in the 1mV range it might be better to spend more effort amplifying rather than moving to higher resolution ADCs. If the real issues are going to be noise and drift, then those might be easier to deal with if the level is boosted first. Microphone preamps can be made from a single 2N3904 transistor and placed in front of a (200x) LM386 modules for less than 50¢ though I suspect there might be lots of distortion. A general purpose (100x) LM358 might do the job on its own, or a (1000x) INA333 or AD623 modules (with the trimpot) which can usually be had for less than $6, as can the AD8221AR. The INA129-HT gets you to 10,000x for ~$9.  What I’d really like is an amplifier with the same simplicity of the ADS1115’s PGA. If anyone knows of a cheap I2C/register controlled opamp module in the hobby market price range, I’d love to hear about it.

Addendum 2020-05-24:   Interrupt latency with wake from sleep

I just watched an interesting video about the sleeping the ESP32 processors and was quite surprised to find out how long (150 µS) and how variable the hardware interrupt latency is on these expressive processors. This set me down the rabbit hole to find out what the latency is on the AVR processors. On a normally running processor you enter the ISR in 23 clock cycles, which is about 1.5µS @16MHz. However if you loop through POWER_DOWN there are extra things to consider like the fact that disabling the BOD in software (just before sleep) is going to add 60 µS to your wake-up time. You also have an ‘oscillator stabilization period’ of 16k CPU cycles with a  standard external oscillator. [see Sect.10.2 of the datasheet] The net result is that the Wake/Start-up time for a 8MHz Arduino is ~1.95ms.  AVR’s with 16MHz clocks like the one I used for this test should have a wake-up time of less than 1ms.  So I was actually cutting it close to combine full POWER_DOWN sleep & the ADS1115’s highest sampling rate. A 3.3v Pro Mini based build @8MHz would not have kept up unless I used SLEEP_MODE_IDLE to keep the main oscillator running which avoids that long stabilization delay.

Other projects using this ADC:

While I’m giving it a B rating for my current use case, this $1 ADC module is probably one of the best options once your signals get above above 10 mv.  The UNO/ADS1115 combo is a ready replacement for benchtop DAQs. Especially since you can add up to four of the modules on the same bus for a multi channel capability.  This build of InstESRE’s Pyranometer solders a PDB-C139 directly onto the ADS1115 module, and adds an analog TMP36 for temperature correction.

If you actually want mains signals in your data, then Open energy monitor has a project reading AC with the YHDC SCT-013-000 . Current sensors like that often make readings that are not referenced to ground so you have to use an ADC capable of differential readings. Although this project focuses getting the most out of cheap eBay modules, the ADS1115 repeatedly makes appearances alongside more pricey sensors like this DIY nitrox tester, and this rather impressive Air Quality Index (AQI) monitor from Ryan Kinnett; Those low power modules from Spec-Sensor look very interesting…

Our LED sensor experiments lead to an interesting observation: When these ‘light-sensing’ loggers are left running overnight they still produce readings because reverse-bias ‘leakage-current’ eventually triggers the Interrupt Capture Unit (ICU) – in the absence of any light. The speed of this self-discharge depends on the ambient temperature. If you cover an rgb LED with black heat shrink, the different color channels have different rates of thermal decay:

Temp (Celsius) vs ‘Covered’ LED reverse-bias discharge time (Seconds) , Red, Blue & Green channels, generic rgb LED.  The LED was encapsulated in black heat shrink tubing and was connected directly to the IO pins with no limit resistor. LED’s take a very long time to discharge compared to other diodes, so at those time scales you can capture the data with the non-ICU based timing method of simply reading the high/low pin state in a loop. We use that simpler method in the 2019 classroom logger starter code on Github.

Both voltage and temperature affect reverse current, so these measurements must start from from a stable, regulated voltage.  Increasing the temperature by almost 20°C reduced the time to 20% of the low-temp value. The green channel appears to be more resistant to leakage which is surprising given that reverse bias currents are usually rated at ~1 µA for RY and ~10 µA for BGW colors. So perhaps this result says more about the volume & surface area of this particular unit than it does about LED color chemistry.

Even if I sleep the processor to save power, multi-minute readings would interfere with the other things we are recording on the Cave Pearl loggers.  However, this LED-based approach has interesting applications where space is limited and temps fall within a warmer range. The idea also has a lot of potential in situations that require high levels of sensitivity, although there aren’t many of those that can wait such a long time for readings.

Checking if I could use the technique with other types of diodes led to this jeelabs post where he compares the reverse bias leakage in three common diodes at 5V:

1N4004 – a high power diode: = 1.3 nA
1N4148 – a low power diode: = 3.4 nA
BAT34 – a Schottky diode:  = 50 nA

He also had the realization that “the reverse current could even be used as a temperature sensor.”  Small diodes have internal capacitance of a few pico-farads, so 5-50 nA will discharge them considerably faster than the LED channels I was using. In fact, reverse leakage increases so much with Schottky diodes it can cause a thermal instability issues which limit their useful reverse voltage to well below their max rating.  Germanium Diodes are even more susceptible.

Add black heat-shrink around  diodes with clear encapsulation (like the one shown here) or they will also be affected by light levels in the local environment.

It’s worth noting that most diode based temperature sensors use the change in forward voltage  because that relationship is linear, with about 2mV less voltage drop for every degree increase of temperature.  But chasing a few milli-volts with Arduino’s 10-bit ADC only allows a precision of ±1°C unless you add amplification, or some other trick.
By comparison, leakage current can be expected to double with every 10°C increase in temperature, making higher resolutions possible with the same hardware. The trade off is using a non-linear relationship which produces variable resolution over the sensing range. And since leakage is also a byproduct of manufacturing variations you need to calibrate each diode individually. That’s a show-stopper in production environments where that time costs more than the whole device, but not so much for DIY projects which need to run-test their build for a few days anyway. We don’t usually send a logger into the field until it’s had several weeks of stable operation.

Testing a 1n5819 Schottky Diode:

Here I’m timing the leakage-discharge with Timer1 clock ticks from an 8mHz 3.3v ProMini:

Temperature (°C) vs 1n5819 Shottky Reverse-bias Leakage Discharge Time (8 mHz clock cycles)  Diode connected between D7 & ICU on D8. Blue dots are Excel’s trend-line. That fit was better than I was expecting.

The Schottky discharges very quickly at room temperatures, with raw Timer1 counts of about 1300 at room temperature ( ~0.16 milliseconds) and about 100 counts of variation /°C.  Counts increase as temperature falls to ~5800 ( ~0.7 ms) at 6°C, with a delta of 580 counts per degree. The curve flattens out at the lower limit of this test with raw counts about 62,000 ( ~7.7 ms) at -15°C, and a delta of 7000 counts/degree.  

The timing jitter on these ICU readings ranges between 10-20 counts depending on the board  (even with 4x noise reduction enabled) and this is a significant source of error when you only have a per-degree delta of 100 counts.  You can over-sample the Schottky to compensate, and testing showed that 256x OS readings produced results that looked very comparable to 1n1418 diodes. (although some authorities say that timing-jitter may be resistant to this smoothing technique.)  Even with oversampling, these short discharge times could become too brief to even count with an 8mhz Promini at temps above 50°C.  However measuring cold temperatures can sometimes be more challenging than warm ones, and for those applications a fast discharging diode like a Schottky might be preferred. With communications overhead, it’s not unusual for an I2C sensor reading to take 1-2ms, so a Schottky might also be better for low power systems trying to minimize CPU runtime.

Testing a 1n4148 Signal Diode:

Temperature (°C) vs 1n4148 Reverse-bias Leakage Discharge Time (8 mHz clock cycles) Diode connected between D7 & ICU on D8. Diode wrapped in black heat shrink tubing. Blue dots are Excel trend-line fit.

The 1n1418 discharges more slowly, with raw Timer1 counts of about 36,000 at room temperatures (~5 msec.), and about 2000 counts of variation/degree at 25°C.  Raw counts increase to ~158000 (~20 ms) at 6°C, with a delta of ~17000 counts per degree and the lower limit of this test saw raw counts 1.3 million (166 ms) at -15°C, and a delta of 140,000 counts/degree.

The 1n1418 is better sensor overall because it won’t drop below the Arduino’s timing capability at natural environment temperatures, and it’s discharge takes long enough that jitter becomes an insignificant source of error. Even in colder environments, 166ms of SLEEP_MODE_IDLE (which leaves Timer 1 running for the clock cycle count) only burns about 0.16 milliamp-seconds per reading on a Promini. That’s not going to break our power budget.

Calibration:

Its worth noting again that you must use a regulated system. Ideally, shifting supply voltage causes a corresponding change to the Schmitt trigger points on the I/O pins. That compensates to some extent, however batteries have significant thermal mass and this causes serious hysteresis problems when sensing temperature.

To calibrate my diodes, I covered them with black heat shrink tubing and taped them in physical contact with an si7051 sensor. Then I placed the logger into a rice filled double ceramic pot ( to add thermal mass) and moved the pot around the kitchen, from the radiators to the refrigerator & freezer. You want stable periods that let the ref & diode sensors equalize, using an average of 20 readings to smooth compressor wobble at the lower end, and those crest/peaks at higher temperatures.

Typical SI7051 (±0.1°C) reference temperature run for calibrating the 1n1418 diode. Boxes indicate plateaus chosen for the calibration data points & coverage areas of closeup graphs shown in the ‘sets’ comparison below.

Excel trend-lines got reasonably close to the response from the Shottky & 1n1418; perhaps needing only one more term for a better fit.  Since thermistors are also semiconductor devices I wondered if those diode decays would generate workable S&H constants if I treat the raw Timer1 counts AS IF they were resistance values from an NTC:

Here I used 20 reading averages to compensate for the fact that the diode is higher resolution than the Si7051 reference, and the long-integration 1n1418  readings have considerably less jitter than the IC sensor.

Then you can convert the discharge time to temperature with the Steinhart-Hart equation:

#define COEFF_A   2.0007E-03    // Coefficient from SRS online calculator
#define COEFF_B   1.3166E-04
#define COEFF_C   3.5441E-09

float Temp = log(RawDiodeDischargeTime);  // note that Log(x) on Arduino is actually LN(x)!
Temp = COEFF_A + (COEFF_B * Temp) + (COEFF_C*(Temp*Temp*Temp));
Temp = (1 /Temp) -273.15;       // -273 converts  Kelvin to Celsius
(Note: I usually save raw readings on the loggers & convert them later in Excel. I’ve been burned several times by loss of significant figures during calculation on the 8-bit 328P processor)

That equation has a quoted accuracy of about ±0.1°C over a 100 degree range when used for a thermistor, but does this hold with a diode sensor?  Yes  – but over a smaller 40 degree range:  (Click Image to Enlarge)

Comparison of si7051 reference temps (blue) (°C) vs 1n1418 based S&H calculations (red) Two examples shown with different ‘center’ points (21°C on the left & 5°C on right) used to generate the three equation constants.

I choose these sets to show calculation errors creeping in as you move farther from the points used to generate the constants. The calculated temperatures in this example drift ~0.07°C from the reference at a distance of ~15°C from the center point.  A tighter set with calibration points at 5, 21, & 36°C produces a near-perfect fit inside that range, with the trade-off  that temps down at -14°C then show an increased deviation >0.1°C.  Overall, it’s about 30% more error than I’d expect to see when calibrating a cheap 10K thermistors with the same points. Given that our Si7051 reference thermometer has a rated accuracy of  ±0.13 °C (datasheet pg 7), I think the best we can achieve for this diode based method is ~±0.2 °C at typical cave temperatures.

So max-middle-min gives you about 40 degrees of usable range and you want at least one of your cal. points at the area of interest.  That’s pretty good considering we are applying the thermistor equation to a different physical system. I will experiment with solver to see if models with more parameters provide a better fit, but this is already is good enough for most of our logger deployments. 

Figure 14-1 I/O Pin Equivalent Schematic from the 328p datasheet. Those protection diodes can also cause problems when de-powering voltage dividers.

My gut feeling is that the re-purposed equation  would work over a wider range if this was a single diode system. However AVR inputs are also connected to two protection diodes and a pull-up MOSFET. Each of these is subject to its own reverse bias leakage to some extent, with the upper protection diode acting in direct opposition to the discharge of the ‘sensor’ diode.  In fact, you can simply run the ICU timing code with nothing at all connected to the D8 pin, and it will still give you a temperature based reading. That makes this the second ‘no parts temperature sensor’  method I’ve discovered for Arduino but, like LED’s, these diodes are low leakage; taking five seconds for a read at 20°C, and five minutes for a reading down at -14°C.  Unless you change the prescaler, the raw numbers could exceed the range of easy calculation on a 328, and show significant hysteresis due to the mass of the chip & Promini board it’s attached to.

The implication here is that temperature sensing via this reverse bias decay method has a sweet spot somewhere between the too-rapid response of a Shottky diode (which approaches the counting limits of an 8MHz clock) and interference from the other stuff connected to Arduino I/O pins. 1n1418’s work well, but I’m sure there are other diodes out there that could do a better job. I have yet to find any good data on the long term stability of reverse bias leakage but we are not stressing the part by exceeding it’s reverse voltage rating, or running enough current to cause much self-heating.  So I suspect that diode leakage is at least as stable as thermistor response over time. There’s a lot of further experimentation to do here, and given the tighter manufacturing spec, I’m curious to see if the method works with diode connected transistors which could make interchangeable temperature sensors possible.

I should also mention that some ‘better quality’ Arduino boards have temperature compensation embedded in their system oscillator. This is a bad thing for this ICU timing method because it introduces a sharp discontinuity in the clock speed when the comp. circuitry kicks in. The S&H constants can’t absorb that like the normal ‘thermal response’ of a cheap oscillator, so the method works better on some boards than others. Another potential problem is moisture accumulating on surfaces -which could provide an alternate current path to discharge the diode. So as with our LED light sensing, desiccants are required inside the logger housing.

the CODE:

D7 is simply acting as a convenient GND connection.

I’ve left this till last, because it’s essentially just a tweaked version of the ICU timing method I posted for reading thermistors. With the diode discharge you triggering on fall instead of rise, and you don’t have to read a reference resistor because we are treating the decay time as a resistance.   The diode’s tiny internal capacitance charges through the INPUT_PULLUP resistor in a few nanoseconds, and there’s no need to discharge afterward.

(Note: This code is modified from the ICU capacitor reading example by Nick Gammon)

(Note: Integer arithmetic on the Arduino defaults to 16 bit & never promotes to higher bit calculations, unless you cast one of the numbers to a high-bit integer first. After casting the Arduino supports 64-bit “long long” int64_t & uint64_t integers for large number calculations but they do gobble up lots of program memory space – typically adding 1 to 3k to the compiled size. Also Arduino’s printing function can not handle 64 bit numbers, so you have to slice them into smaller pieces before using any .print functions) 

Addendum 2021-01-24        Don’t sleep with regular I/O pins.

I’ve been noodling around with other discharge timing methods and come across something  that’s relevant to using these methods on other digital pins. Here’s the schematic from the actual 328 datasheet, with a bit of highlighting added. The green path is PINx.  It’s always available to the databus through the synchronizer (except for in SLEEP mode?) The purple path is PORTx.   Whether or not it is connected to PINx depends on the state of DDRx (which is the yellow path.)

As shown in the figure of General Digital I/O, the digital input signal can be clamped to ground at the input of the Schmitt Trigger. The signal denoted SLEEP in the figure, is set by the MCU Sleep Controller in Power-down mode and Standby mode to avoid high power consumption if some input signals are left floating, or have an analog signal level close to VCC/2.

When sleeping, any GPIO that is not used an an interrupt input has its input buffer disconnected from the pin and in clamped LOW by the MOSFET.

Clearly D8 on the ICU must one of those ‘interrupt exceptions’ or the thermal discharge of the diode would have been grounded out by entering the sleep state.  If you use a similar method on regular IO pins you can’t sleep the processor in that central do-while loop.